Phase-shifted full-bridge topology with current injection

ABSTRACT

Systems and method for optimizing the efficiency of operation of electronic circuits, configured structured according to a true soft-switching phase-shifted full-bridge topology (where all the primary switching elements turn on at zero voltage and the secondary switching elements turn off at zero current with no ringing and no spikes across the secondary switching elements) with the use of unique current-injection approaches. An additional advantage of the embodiments of this invention is that the true soft switching feature applies regardless of the leakage inductance in the transformer.

CROSS-REFERENCE TO RELATED APPLICATIONS

This patent application is a continuation in part from the U.S. patentapplication Ser. No. 15/987,499 filed on May 23, 2018 and now publishedas US 2018/0278169, which, in turn, is a continuation-in-part from theU.S. patent application Ser. No. 15/068,598 filed on Mar. 13, 2016, nowpublished as US 2017/0012547, which claims priority from the U.S.Provisional Patent Application No. 62/133,245 filed on Mar. 13, 2015.

The disclosure of each of the above-identified patent applications isincorporated herein by reference.

TECHNICAL FIELD

The invention relates to power converters configured according to a truesoft switching phase-shifted full-bridge topology and, in particular, tosuch transformers in which the true soft switching occurs regardless ofthe value of the leakage inductance in a converter.

Over the years, the term “soft switching technologies” has been used dorefer to technologies in which the primary switching elements(interchangeably referred to as primary switcher) in a converter areturned on at zero voltage. These technologies, however, are notconfigured to create soft switching across the secondary switchingelements (interchangeably referred to as secondary switchers). Softswitching for the secondary switching elements of a power converterimplies that the secondary switching elements turn off at zero currentand there is substantially or completely no ringing and/or spikes ofvoltage across the secondary switching elements at time of turn off.Elimination of the ringing and spikes across the switching elements inthe secondary implies that the turn off is carried out when the currentthrough the secondary switching elements reaches zero or a smallnegative level and that the charge of the parasitic capacitance acrossthe secondary switching elements is effectuates with a current source.

One of the most popular soft switching topologies is the phase-shiftedfull-bridge topology. Thus far, in order to obtain soft switchingconditions across the primary switching elements, considered the leakageinductance of the converter to be a very important parameter, and suchleakage inductance in the transformers was intentionally increased. Intraditional soft switching phase-shifted full-bridge topologies, zerovoltage switching was obtained by using the energy in the leakageinductance to discharge the parasitic capacitance reflected across theprimary switches (˜reflected through the transformer in the primaryside). Sometimes, additional inductive elements were used as elementsplaced in series with the primary winding of the transformer to create avirtual leakage inductance. In some cases, the magnetizing currentamplitude was increased in the transformer or in an inductor to form avirtual magnetizing current, in order to have enough energy inmagnetizing current to discharge the parasitic capacitances across theprimary switches.

Phase-shifted full-bridge topologies known to date do not ensure softswitching across the secondary rectifiers while the converter isoperating in continuous mode. (As a result, large voltage spikes andringing typically occur across the secondary rectifiers, whichnegatively affect the efficiency of the overall circuitry and thesuppression of which requires the use of an additional electricalapparatus configured to be effective in protecting the transformers fromelectrical transients (known as a snubber).

SUMMARY

Embodiments of the invention provide a specifically-designed electroniccircuitry for a DC-DC converter and methods for operation of same. Inparticular, embodiments of the invention provide a method for operatinga pulse-shifted full-bridge (PSFB) DC-DC converter that includes: aprimary side and a secondary side; a transformer having at least oneprimary winding at the primary side and at least one secondary windingat the secondary side, wherein a leakage inductance is formed betweenthe at least one primary winding and at least one secondary winding; abridge formed by two legs connected in parallel at the primary side, oneleg being a linear leg and another leg being a resonant leg. Here, eachleg is formed by corresponding bottom primary switching element andupper switching element at the primary side configured in a totem polearrangement; where common terminals of the two legs are connected to aninput voltage source; where shared terminals of switching elementswithin one leg, from the two legs, are connected to one end of at leastone primary winding and wherein shared terminals of the switchingelements of another leg, from the two legs, are connected to another endof at least one primary winding; where primary switching elements of agiven leg, from the two legs, are configured to be complementarty toeach other during operation of the converter with a period of dead timethat includes driving signals from one leg to be phase-shifted withrespect to driving signals from another leg. The converter furtherincludes first and second synchronous rectifiers at the secondary side;at least one output inductor at the secondary side, and acurrent-injection electronic circuit. In the at least one outputinductor at the secondary side, a first terminal of the at least oneoutput inductor is connected to a load of the converter, while a secondterminal of the at least one output inductor is directly connected to asynchronized rectifier from the first and second synchronous rectifiers.In the current-injection electronic circuit, there are a) twocurrent-injection switching elements, respectively corresponding toswitching elements in the resonant leg as well as b) twocurrent-injection windings disposed on the secondary side and coupled tothe at least one secondary winding of the transformer; c) twocurrent-injection capacitors; two diodes; and d) a voltage-injectionvoltage source. Here, the two current-injection switching elements areconnected to respectively-corresponding first terminals of twocurrent-injection windings; and each of respectively-correspondingsecond terminals of the two current-injection windings is connected to acorresponding current-injection capacitor from the two current-injectioncapacitors. Furthermore, a cathode of each of the two diodes isconnected to the corresponding current-injection capacitor at thecorresponding second terminal and an anode of each of the two diodes isconnected to the voltage-injection voltage source. Such converteroperates according to the following operational steps:

1) switching on an upper primary switching element of the resonant legand a bottom primary switching element of the linear leg, where upperprimary switching element of the resonant leg and the bottom primaryswitching element of the linear leg defines a first diagonal of thebridge and, while the first synchronous rectifier is on, transferringpower from the primary side to the secondary side (such transferringbeing characterized by linearly changing, with time, a first amplitudeof first current flowing through the at least one output inductor andlinearly increasing a second amplitude of magnetizing current of thetransformer to a peak value of the second amplitude). 2) After switchingoff the bottom primary switching element of the linear leg and turningon the upper primary switching element of the linear leg, continuing thetransferring power to the load and continuing the linearly changing ofthe amplitude, of current flowing through the at least one outputinductor, to a lowest value of said first amplitude while maintainingthe second amplitude of the magnetizing current at the peak value. 3)After switching off the upper switching element of the resonant leg,turning on a current-injection switching element corresponding to abottom switching element of the resonant leg with a time delay withrespect to the moment of said switching off the upper switching elementto reflect an injection current, flowing through said current-injectionswitching element, in a secondary winding.

A related method for operation of the above-described converter (furthercomplemented with a controlling electronic circuitry configured togenerate control signals to the primary switching elements, the controlsignals having square waveforms), includes the following steps:

(1) Switching on an upper primary switching element of the resonant legand a bottom primary switching element of the linear leg, where theupper primary switching element of the resonant leg and the bottomprimary switching element of the linear leg define a first diagonal ofthe bridge and, while the first synchronous rectifier is on,transferring power from the primary side to the secondary side. Suchtransferring is characterized by linearly changing, with time, a firstamplitude of first current flowing through the at least one outputinductor and linearly increasing a second amplitude of magnetizingcurrent of the transformer to a peak value of the second amplitude. (2)After switching off the bottom primary switching element of the linearleg and turning on the upper primary switching element of the linearleg, continuing the transferring power to the load and continuing thelinearly changing of the amplitude (of current flowing through the atleast one output inductor) to a lowest value of the first amplitudewhile maintaining the second amplitude of the magnetizing current at thepeak value. (3) After switching off the upper switching element of theresonant leg, discharging a parasitic capacitance reflected acrossprimary switching elements of the resonant leg, with the use of theleakage inductance. (4) Before switching off the upper switching elementof the resonant leg, turning on a current-injection switching elementcorresponding to a bottom switching element of the resonant leg with atime interval that is adjustable with respect to the moment of switchingoff the upper switching element to reflect an injection current (flowingthrough said current-injection switching element) in a secondarywinding. (5) Switching off the first synchronous rectifier after a sumof a magnetizing current (reflected in the secondary winding) and theinjection current (reflected into the secondary winding) exceeds thelowest value of the first amplitude by a predefined excess amount ofcurrent to cause the excess amount of current to flow into at least oneprimary winding while continuing said discharging the parasiticcapacitance towards zero.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 schematically illustrates an electronic circuitry of DC-DC powerconverter configured according to a phase-shifted full-bridge topologywith center tap secondary side and current injection.

FIG. 2 schematically illustrates the key waveforms representing theoperation of the embodiment of converter of FIG. 1.

FIG. 3 depicts a section of the current injection circuit of theembodiment of FIG. 1.

FIG. 4 contains plots illustrating time-dependency of current passingthrough the diode of the portion of the current injection circuit ofFIG. 3 as a function of the amplitude of the voltage provided by thevoltage source of the current injection circuit.

FIG. 5 illustrates the injection current, delivered by the portion ofthe current injection circuit of FIG. 3, and the current passing throughthe additional diode of the current injection circuit.

FIG. 6 depicts the control signals for the primary switching elementsand the timing of the identified injection current according to anembodiment of this invention.

FIG. 7 shows plots representing several key waveforms associated withthe current injection and the conditions at which the current injectionreaches the peak value.

FIG. 8 addresses the most desirable timing for reaching the peak of thecurrent injection into the embodiment of the converter as a function ofvoltage at the switching node in the primary side of the converter.

FIGS. 9A, 9B, 9C presents methodology of obtaining the current injectionvoltage source for the current injection circuit.

FIG. 13 shows the electronic circuitry structured according to thefull-bridge-with-current-injection topology and configured to transferthe energy in reverse (from the low-voltage buss to the high-voltagebuss).

FIG. 14 shows the key waveforms for the circuit of FIG. 13.

FIG. 15 illustrates the key waveforms from the circuit from FIG. 13 withemphasis on the active clamp circuit.

FIG. 16 contains plots representing the key waveforms of the activeclamp circuit as part of the circuit depicted in FIG. 13.

Each of the plots representing a key waveform shows a dependency ofparticular characteristic(s) or parameter(s) as a function of time.Generally, the sizes and relative scales of elements in Drawings may beset to be different from actual ones to appropriately facilitatesimplicity, clarity, and understanding of the Drawings. For the samereason, not all elements present in one Drawing may necessarily be shownin another.

DETAILED DESCRIPTION

In the embodiments of this invention, through different means thecurrent through the synchronous rectifiers is caused to reach zerobefore a switching element in a resonant leg of the circuitry turns on.At the moment when one of the synchronous reciters turn off and themagnetizing current can provide the current to the output inductor, andthe additional current above the current demanded by the output inductorwill flow into the primary discharging the parasitic capacitances of theprimary switching elements to zero. Unlike the traditional method ofobtaining soft switching using the energy in the leakage inductance inthis technology zero voltage switching is guaranteed at and for anyloading condition. To achieve the present goal, a process of dischargeof the parasitic capacitances across the switching elements from a partof the resonant leg (by the magnetizing current and current from currentinjection source(s) introduced to the circuitry) starts to after theentire, total energy in the leakage inductance is used, and the voltageacross the primary switching elements reach the specific lower level(the leakage current energy discharges the primary switching elements ofsuch lower level, after which the magnetizing current and the currentinjection take off). The proposed methodology is operable with andapplicable to any strength (value) of leakage inductance (including asituation in which the leakage inductance is absent) once the summationof the magnetizing current and the injection current is at a certainpredetermined level, thereby eliminating an existing demand to have aminimum specified leakage inductance in the transformer.

Example 1

FIG. 1 is a simplified schematic of a power converter structuredaccording to a full-bridge topology with the center tap secondary side,which utilizes synchronous rectifiers as rectification means and that isconfigured to use current injection to obtain zero-voltage switching onor for all of the present switching elements in any operating condition.The electronic circuitry configured according to the soft-switchingfull-bridge topology is composed by two half-bridge legs, one formed bythe combination of the switches (M1,102) and (M2, 104) and another—bythe combination of the switched or switching elements (M3, 106) and(M4,108). In the full-bridge topology, the switches M1 and M2 form theso-called linear leg, and the switches M3 and M4 form the so-calledresonant leg. The switching elements from the linear leg turn on at zerovoltage switching conditions regardless of operating condition of theconverter. When any of the switching elements within the linear leg isturned on, the voltage induced in the secondary winding is substantiallyzero. When any of the switching elements of the resonant legs is turnedon, the voltage induced in the secondary winding is substantially largerthan zero. The primary side of the transformer (Tr1, 130), representedby the primary winding (L1, 132), is connected to the full-bridgeelectronic structure through the terminations or termination contacts(TA, 200) and (TB, 202). The secondary windings of the transformer (Tr1,130) are denoted as (L2, 134) and (L3, 136) and are connected to thesynchronous rectifiers (SR_(A), 118) and (SR_(B), 120). The secondarywindings are also connected to the output inductor (Lo, 126) and outputcapacitor (Co, 128). The voltage across the output capacitor (Co, 128)is denoted as (Vo, 168).

Besides the key electronic components that form the full-bridge topologyportion of the overall circuitry of FIG. 1, this overall circuitryincludes the current injection circuit 164 (outlined with a dashedline), the use of which is described below.

The current injection electronic circuit 164 contains two sub-circuits(one associated with the auxiliary winding (L4, 138) and thecorresponding auxiliary switch (Minj1, 142), and another—with theauxiliary winding (L5, 140) and the corresponding auxiliary switch(Minj, 144)). The two sub-circuits in operation produce currents (Iinj1,150) and (Iinj2, 152), as discussed below. These two currents areactivated by two control signals, (Vcinj1, 146) and (Vcinj2, 148)applied to the switches (Minj, 142) and (Minj2, 144)respectively-corresponding to the sub-circuits at hand.

The overall electronic circuitry 1000 of FIG. 1 additionally includesthe controller 999, which is appropriately configured to produce thesignals governing the operation of the primary switches in the primaryside, for the synchronous rectifiers in the secondary side, and also forthe current injection circuit 164.

As a person of skill in the art will readily appreciate, the electroniccircuitry of FIG. 1 can be transformed to a circuitry structuredaccording to a phase-shifted full-bridge (PSFB) topology, if the controlsignals for the primary switching elements, M1, M2, M3 and M4 arejudiciously timed as depicted in FIG. 2.

The switching elements (M1,102) and (M2, 104) of FIG. 1, which togetherform the linear leg in the primary side of the converter, areoperationally-complementary to each other, as depicted in FIG. 2. Whenone of these switching devices is “on”, the other is “off”. There is asmall time interval between the period of “on” operation of theseswitches, when both switching devices are “off”, referred to as a “deadtime” interval, DT. The switching elements or devices (M3, 106) and (M4,108) of the resonant leg are also operationally-complementary to eachother, as depicted in FIG. 2.

In operation of the converter of FIG. 1, the energy is transferred fromthe primary winding (L1, 132) to the secondary windings (L2,134) and(L3, 136) only when switches in the pair of switches (M1, 102) and (M4,108) conduct (are “on”) at the same time or switches in the pair ofswitches (M3, 106) and (M2, 104) conduct (are “on”) at the same time.

The switching elements M1 and M2, placed in the linear leg, turn on atzero voltage switching conditions under most of the operatingconditions. The switching elements M3 and M4 do not turn “on” at zerovoltage switching conditions except if special requirements are met.Various solutions have been proposed of how to ensure that the switchingelements located on the resonant leg can be turned “on” at zero voltageswitching conditions, including those discussed in U.S. Pat. Nos.5,231,563; 6,862,195; 7,009,850; and 9,985,546 (all of which,aggregately or individually, are referred to as Our Prior Publications).U.S. Pat. No. 9,985,546, for example, discloses a method for obtainingzero voltage switching conditions in the phase-shifted full-bridgetopology by utilizing the magnetizing current in addition to a currentinjection in the transformer. The disclosure of the present invention isdirected to improving the efficiency of the current injectionmethodologies for use in power converters.

FIG. 2 contains plots illustrating the key waveforms of the operation ofthe PSFB topology equipped with current injection. Time moments labelledas t_(CRA) and t_(CRB) identify critical times when switching devicesfrom the resonant leg of FIG. 1 are turned “on”. Specifically, t_(CRA)is the time when the switching element (M3, 106) turns “on”, and t_(CRB)is the time when the switching element (M4, 108) turns “on”. At thesetimes, the voltage across the switching elements that turn “on” is setto zero in order to obtain zero voltage switching conditions.

As depicted in FIG. 2, the injection current Iinj1 is used to ensurethat the switch (M3, 106) is turned “on” at zero voltage switchingconditions by discharging the parasitic capacitance reflected across theswitch M3. The injection current Iinj2 is used to ensure that the switch(M4, 108) turns “on” at zero voltage switching conditions by dischargingthe parasitic capacitance reflected across such switch.

FIG. 3, which presents only a portion or section of the currentinjection circuit 164 of the embodiment 1000 of FIG. 1, facilitates theunderstanding of the embodiment of FIG. 1. The shown portion 3000 of thecurrent injection circuit 164 is composed of the auxiliarycurrent-injection winding (L4, 138); the auxiliary switch (Minj1, 142),controlled by a control signal (Vcinj1, 146): and a current injectioncapacitor (Cinj1, 154). The U.S. patent application Ser. No. 15/987,499,the disclosure of which is incorporated herein by reference, disclosed acurrent injection circuit formed by a current injection switch, acurrent injection winding, and a current injection capacitor. As wasdiscussed in U.S. patent application Ser. No. 15/987,499, for thatcurrent injection circuit of the related art to work in a way whereinthe current injection can be controlled by the phase shift between thecontrol signal of the current injection and the control signal of thecorresponding primary switch, the capacitance of the current injectioncapacitor Cinj1 has to have such a high value that the voltage rippleacross this capacitor is much smaller than the average voltage value.Such solution, presented in the U.S. patent application Ser. No.15/987,499, leads to higher RMS current through the current injectionswitch and the current injection winding, because the energy provided bythe current injection capacitor to the current injection has to bereplenish at each cycle of the operation, thereby leading to circulationof large energy and, as a result, an increase of the RMS current.

In advantageous distinction from the current injection circuit of U.S.patent application Ser. No. 15/987,499, the embodiment of the currentinjection circuit according to the idea of the present invention, isequipped with an additional diode (Dinj1,158) and a voltage source(Vinj1,162). Depending on the specifics of the particularimplementation, the voltage source (Vinj, 162) can be configured tooperate in a varying fashion, for example such as that when the voltageis controlled by a programmable processor according to a specificalgorithm.

As the skilled person will readily appreciate, the portion of thecurrent injection circuit 164 implemented according to the embodiment3000 offers substantial flexibility to the electronic-circuit designer.By sizing the value of the capacitance of (Cinj1, 154), one can controlthe resonant energy that is delivered by (the (Cinj1, 154), as well asthe energy coming from (Vinj, 162) via the diode (Dinj1, 158). As aresult, the shape of the current injection pulse delivered to thecircuitry can also be controlled. The purpose for controlling the shapeof the current injection pulse is to minimize the RMS current of thecurrent injection (which remained an operational shortcoming in U.S.patent application Ser. No. 15/987,499) and to be able to discharge theparasitic capacitance of the switching element on the primary sidecorresponding to the particular injection current. For example, in FIG.1, the primary switching element M3 corresponds to the injection current(Iinj1, 150) produced by the circuit 164, while the primary switchingelement M4 corresponds to the injection current (Iinj2, 152) produced bythe current injection circuit 164.

FIG. 4 contains plots illustrating time-dependency of current passingthrough the diode (Dinj, 158) as a function of the amplitude of Vinj.The quantity (VinjN, 170) represents the voltage in the currentinjection winding (here—in L4) when the corresponding current injectionswitch (here—(Minj1,142)) is “on”. It can be deduced from FIG. 4 thatthe best voltage level for Vinj is VinjN.

FIG. 5 illustrates the injection current Iinj1, delivered by the portion3000 of the circuit 164, and the current through the Dinj1. Inoperation, the injection current Iinj1 is the summation (or aggregate)of the current through Dinj1 and the resonant current through Cinj1. Theportion of the Iinj 1 cycle where the injection current is shown to havenegative amplitude (that is, the time interval between t1 and t4) istailored by the capacitance value of the Cinj1. This portion of the Iinj1 cycle (the negative injection current between times t1 and t4) is veryimportant for the proper operation of the current injection circuit.Indeed, without the presence of the negative portion of the injectioncurrent, the current injection switch Minj1 would turn “off” when apositive current flows through it, thereby leading to very high voltagespikes across the switch Minj1. According to the idea of the invention,however, the current injection switch (Minj1, 142) is turned “off” atthe moment t3, which is a time moment chosen between t1 and t4 while theIinj1 current is negative. The current Iinj1 continues to flow throughMinj1 after Minj1 turns “off”.

During the operation of the current injection circuit 164, the capacitorCinj1 plays the role of shaping the time-dependent profile of theinjection current, Iinj1, to create the required negative current and toincrease the injection current amplitude as needed. The goal for theoptimum shape of the injection current is to have a low RMS value and beable to discharge the parasitic capacitance of the correspondingswitching element.

FIG. 6 contains plots representing time-dependent control signals forthe primary switching elements (shown as having square-shaped waveforms)and the control signals for one of the current injection controlsignals, in this case—Vcinj2, configured according to an embodiment ofthe invention. (The electronic schematic and key waveforms weredescribed in reference to FIGS. 1 and 2). As depicted in FIG. 6, thecontrol signal for Minj2 is activated ahead of and prior to the controlsignal for the primary switch M4, by a time period a. In someapplications, the duration of this lead-time-interval a can be evenlarger than the duration Tdead of the “dead time” interval between theoperation of the primary switches in each of the linear and resonantlegs. As depicted in FIG. 6, the control signal for the currentinjection can overlap with the control signal for the correspondingprimary switch (M3 in this case) with some operational benefits.

FIG. 7 illustrates several key waveforms that play role in shaping theinjected current according to an embodiment of the invention. Thewaveforms presented in FIG. 7 include V(A), where A is the switchingnode of the resonant leg 174 depicted in FIG. 1.

As presented in FIG. 7, at the moment t0—when the primary switch M3turns “off”—the voltage in the switching node A starts decaying towardsthe zero level. The control signal for (Minj2, 144)—that is the signalVcinj2—is turned “on” at the time when M3 is turned “off”. VinjN2 startsbuilding up as the voltage in node A decays. The voltage across Cinj2starts decaying because a portion of the injection current Iinj2 isdelivered from the Cinj2 capacitor. At the moment t_(PK), the voltageacross the capacitor Cinj2 equalizes with the level of VinjN2, when theinjection current Iinj2 reaches its peak value. Iinj peak2, and afterthat moment the injection current Iinj2 starts decaying. The voltagelevel at the node A at the moment when Iinj2 reaches its peak value isshown as Z. In one implementations of the idea of the invention, thecurrent injection is configured to reach its peak at a voltage levelwithin the range from about 0.125*Vin and below about 0.5*Vin.

To that end, a related embodiment of injection current and the voltageat the switching node A of the primary side of the transformer ispresented in FIG. 8: this embodiment represents the optimized situationwhere the Z voltage level (at which injection current reaches its peak)is between about 0.375*Vin and about 0.250*Vin. In an ideal resonantcircuit, where a capacitor is resonantly discharged by a half resonantshape current, the peak current value would be at a Z voltage level ofabout 0.5*Vin. (In comparison, in embodiments disclosed in U.S. patentapplication Ser. No. 15/987,499, where the capacitance of the Cinjcapacitor has a value large enough to ensure that the ripple across thecapacitor Cinj is much smaller than average voltage across the capacitorCinj, the peak value of the injected current is reached at the moment oftime that comes later, after the voltage at switch node A reaches zero,after t_(ZVS), which is the time at which the zero-voltage switchingconditions are achieved). The optimized current injection methodology,configured according to an embodiment of the invention, requires thatthe discharge of the parasitic capacitance reflected across of theprimary switching element be done by current injection with minimum(minimized) RMS value. As can be seen in FIG. 8, after the voltage inthe switch node A reaches zero level at the moment t_(ZVS), the currentinjection amplitude should decrease rapidly to reduce the RMS current ofthe current injection.

As the person of skill will readily appreciate, the circuit for thecurrent injection a portion 3000 of which is presented in FIG. 3 canaccomplish the goal of a low RMS value of the injected current, producedby such circuit, while maintaining the capability of the currentinjection to discharge the parasitic capacitance reflected across thecorresponding primary switch. For example, the parasitic capacitancereflected across M4 is discharged by the injection current Iinj2, andthe parasitic capacitance reflected across M3 is discharged by theinjection current Iinj1, as per FIG. 1. The portion of the currentinjection circuit 3000 depicted in FIG. 3 (in which the energy for thecurrent injection is delivered by i) a voltage source (Vinj, 162)producing voltage of a suitable amplitude close to VinjN and ii) fromthe resonant capacitors (Cinj1, 154) and respectively (Cinj2, 156) canoptimize the shape and the RMS current of the current injection for thehighest efficiency.

Notably, unlike in the methodology disclosed by Mao in U.S. Pat. No.7,548,435 (where the current injection is only resonant and theamplitude of the current injection cannot be controlled through a phaseshift or other means), the embodiment of the invention are configured toenable a full control of the current injection amplitude by a phaseshift between the control signal for the current injection switch andthe control signal of the corresponding primary switch: In reference toFIG. 6, the amplitude of the current injection is controlled bycontrolling the value of α. Additionally—and unlike in the case of thecircuitry discussed in U.S. patent application Ser. No. 15/987,499—thecontrol signal for the current injection switches, Minj1 and Minj2, isallowed to overlap with the control signal of the complementary switchof the corresponding switch on the resonant leg of the circuitry 1000.By turning “on” the current injection switch at the moment t0—that is,ahead of the moment t_(Z1) when the complementary switch of thecorresponding primary switching element is turned “off” (this timeinterval is denoted as δ in FIG. 6), the current injection Iin2 iscaused to build up ahead of time, which speeds up the discharge of theparasitic capacitance. In addition to that, the increase of theamplitudes of the VinjN1 and respectively VinjN2 waveforms is delayed,as depicted in FIG. 7, which delay affects the increase of the rate ofrise of the injection currents Iinj1 and respectively Iinj2. Bydecreasing the time difference between the moments t0 and tpk, the RMScurrent of the current injection is decreased (for the same peak valueof the Iinjpeak).

Referring now to FIGS. 9A, 9B, and 9C, which will be readily understoodby a person of skill in the art, different methods of structuring thevoltage source Vinj are illustrated. In particular, FIG. 9C illustratesthe situation when the voltage injection source is realized by using theoutput voltage of the converter (if that voltage is close in value toVinjN from FIG. 3).

Example 2

Here we discuss an example of a Phase-Shift Full-Bridge topology withcurrent injection configured to operate as a current-fed push-pullconverter. In some applications—such as, for example, an auxiliarybattery charger for automotive industry, a DC-DC converter must have thecapability to transfer energy in reverse: from the low voltage (forexample 12 V battery) to the high voltage, (for example, a 400 V buss).FIG. 13 illustrates an example of a phase-shifted full-bridge topologyof the electronic circuitry, which in operation transfers the energyfrom the low voltage buss to the high voltage buss. Such mode ofoperation of a power converter is referred to herein as a current-fedpush-pull power conversion.

Here, the energy comes from the low voltage side 1300A of thetransformer that has (V_(LV), 500) and is transferred to the highvoltage buss 1300B having (V_(HV), 506). Additionally is shown a portionof the current-injection circuit. FIG. 14 contains plots illustrating,as functions of time, the key waveforms representing the operation ofthe converter circuit of FIG. 13 and including Vc_(SRA) (the controlsignal for synchronous rectifier SR_(A)) Vc_(SRB) (the control signalfor the synchronous rectifier SR_(B)), I_(L2) (representing the currentthrough the winding L2), I_(L3) (representing the current passingthrough the winding L3), and I_(L0) (representing the current passingthrough the output inductor Lo)

The switching elements SR_(A) and SR_(B) are “on” during the same timeduring the time interval between t_(b0) and t_(b1) and during the timeinterval between t_(b2) and t_(b3). The period(s) of simultaneousconduction SR_(A) and SR_(B) is necessary in order to create a boostaction and be able to control the high voltage buss amplitude, V_(HV).Because the coupling in the transformer Tr1, between/among the windingsL1, L2, and L3 it does not reach 100%, there exists a leakage inductancebetween/among L2, L3, and L1. As a result, when the synchronousrectifiers (SR_(A), 118), and (SR_(B), 121) are turned off, the energyof leakage inductance in the transformer Tr1 causes voltage spikesacross the rectifiers SR_(A) and SR_(B).

In FIG. 15 are depicted the key waveforms of the low-voltage section ofthe converter of FIG. 13, which waveforms include the current I_(L0)through the output inductor (Lo, 126); the current I_(L2) through L2;the control signal for SR_(A): Vc_(SRA); the control signal for theswitch M_(C1A) and the current I_(CclA) through the clamp capacitorC_(clA).

As a person of skill will readily appreciate, the active clamp circuitformed by M_(clA) and C_(clA)—and that formed by another pair of theswitch M_(clB) and clamp capacitor C_(clB)—operate as active clamps. Forexample, in operation, when SR_(A) turns “off”, the leakage inductanceexisting between the windings L2 and L3 and L1 will force some of thecurrent flowing through SR_(A) to flow via the switch M_(clA) for ashort period of time until M_(clA) is turned “on”. That current flowwill charge the clamp capacitor C_(clA), which now acts as a currentmirror and push back into the L1 winding and further to the V_(HV)high-voltage buss. The same developments occur when SR_(B) is turned“off”, as a result of which the leakage inductance current flowingthrough L3 is caused to flow through M_(clB) towards C_(clB) and furthermirrored back to the high-voltage buss via the winding L1.

FIG. 16 illustrates the control signal Vc_(SRA) for the synchronousrectifier SR_(A), the control signal Vc_(MclA) for the switch M_(clA),and the current Ic_(lA) flowing through C_(clA) (and reaching zero levelbefore the moment of time when MclA turns “off”). For a lower value ofthe capacitance C_(clA), the shape of the current waveform Ic_(lA) ischanged, and during the period of conductance of the switch M_(ClA)assumes the form depicted by the portion I_(clAx) of the curve. Thecurrent through the clamp capacitor will be shaped as depicted byI_(ClAX) when the leakage inductance energy between L2 and L1 istransferred to V_(HV) buss 1300B before the clamp switch M_(clA) turns“off”.

It is appreciated that described methodologies can also be applied tooperation of other topologies (such as asymmetrical half bridges andasymmetrical full bridges, push pull, half bridges, and single ended anddouble ended forward topologies, as well as to any other derivation offorward derived topologies; center tap, and full bridge rectificationcircuitries).

For the purposes of this disclosure and the appended claims, the use ofthe terms “substantially”, “approximately”, “about” and similar terms inreference to a descriptor of a value, element, property orcharacteristic at hand is intended to emphasize that the value, element,property, or characteristic referred to, while not necessarily beingexactly as stated, would nevertheless be considered, for practicalpurposes, as stated by a person of skill in the art. These terms, asapplied to a specified characteristic or quality descriptor means“mostly”, “mainly”, “considerably”, “by and large”, “essentially”, “togreat or significant extent”, “largely but not necessarily wholly thesame” such as to reasonably denote language of approximation anddescribe the specified characteristic or descriptor so that its scopewould be understood by a person of ordinary skill in the art. In onespecific case, the terms “approximately”, “substantially”, and “about”,when used in reference to a numerical value, represent a range of plusor minus 20% with respect to the specified value, more preferably plusor minus 10%, even more preferably plus or minus 5%, most preferablyplus or minus 2% with respect to the specified value. As a non-limitingexample, two values being “substantially equal” to one another impliesthat the difference between the two values may be within the range of+/−20% of the value itself, preferably within the +/−10% range of thevalue itself, more preferably within the range of +/−5% of the valueitself, and even more preferably within the range of +/−2% or less ofthe value itself.

The use of these term in describing a chosen characteristic or conceptneither implies nor provides any basis for indefiniteness and for addinga numerical limitation to the specified characteristic or descriptor. Asunderstood by a skilled artisan, the practical deviation of the exactvalue or characteristic of such value, element, or property from thatstated falls and may vary within a numerical range defined by anexperimental measurement error that is typical when using a measurementmethod accepted in the art for such purposes.

Other specific examples of the meaning of the terms “substantially”,“about”, and/or “approximately” as applied to different practicalsituations may have been provided elsewhere in this disclosure.

An embodiment of the system generally may include electronic circuitry(for example, a computer processor and/or controller) governing anoperation of the embodiment and controlled by instructions stored in amemory, to perform specific data collection/processing and calculationsteps as disclosed above. The memory may be random access memory (RAM),read-only memory (ROM), flash memory or any other memory, or combinationthereof, suitable for storing control software or other instructions anddata. Those skilled in the art should would readily appreciate thatinstructions or programs defining the operation of the presentembodiment(s) may be delivered to a processor in many forms, including,but not limited to, information permanently stored on non-writablestorage media (e.g. read-only memory devices within a computer, such asROM, or devices readable by a computer I/O attachment, such as CD-ROM orDVD disks), information alterably stored on writable storage media (e.g.floppy disks, removable flash memory and hard drives) or informationconveyed to a computer through communication media, including wired orwireless computer networks. In addition, while the invention may beembodied in software, the functions necessary to implement a method ofthe invention may optionally or alternatively be embodied in part or inwhole using firmware and/or hardware components, such as combinatoriallogic, Application Specific Integrated Circuits (ASICs),Field-Programmable Gate Arrays (FPGAs) or other hardware or somecombination of hardware, software and/or firmware components.

The invention as recited in claims appended to this disclosure isintended to be assessed in light of the disclosure as a whole. Variouschanges in the details, steps and components that have been describedmay be made by those skilled in the art within the principles and scopeof the invention.

While the invention is described through the above-described exemplaryembodiments, it will be understood by those of ordinary skill in the artthat modifications to, and variations of, the illustrated embodimentsmay be made without departing from the inventive concepts disclosedherein. Accordingly, the invention should not be viewed as being limitedto the disclosed embodiment(s).

1. A method for operating a pulse-shifted full-bridge (PSFB) DC-DCConverter, the converter comprising a primary side and a secondary side;a transformer having at least one primary winding at the primary sideand at least one secondary winding at the secondary side, wherein aleakage inductance is formed between the at least one primary windingand at least one secondary winding; a bridge formed by two legsconnected in parallel at the primary side, one leg being a linear legand another leg being a resonant leg, wherein each leg is formed bycorresponding bottom primary switching element and upper switchingelement at the primary side configured in a totem pole arrangement,wherein common terminals of the two legs are connected to an inputvoltage source, wherein shared terminals of switching elements withinone leg, from the two legs, are connected to one end of at least oneprimary winding and wherein shared terminals of the switching elementsof another leg, from the two legs, are connected to another end of atleast one primary winding, wherein primary switching elements of a givenleg, from the two legs, are configured to be complementarty to eachother during operation of the converter with a period of dead time thatincludes driving signals from one leg to be phase-shifted with respectto driving signals from another leg; first and second synchronousrectifiers at the secondary side; at least one output inductor at thesecondary side, wherein a first terminal of said at least one outputinductor is connected to a load of the converter, wherein a secondterminal of said at least one output inductor is directly connected to asynchronized rectifier from the first and second synchronous rectifiers;and a current-injection electronic circuit that includes: twocurrent-injection switching elements, respectively corresponding toswitching elements in the resonant leg; two current-injection windingsdisposed on the secondary side and coupled to the at least one secondarywinding of the transformer; and two current-injection capacitors; twodiodes; and a voltage-injection voltage source; wherein said twocurrent-injection switching elements are connected torespectively-corresponding first terminals of two current-injectionwindings, wherein each of respectively-corresponding second terminals ofthe two current-injection windings is connected to a correspondingcurrent-injection capacitor from the two current-injection capacitors;wherein a cathode of each of the two diodes is connected to saidcorresponding current-injection capacitor at the corresponding secondterminal and an anode of each of the two diodes is connected to thevoltage-injection voltage source; the method comprising: (a) Switchingon an upper primary switching element of the resonant leg and a bottomprimary switching element of the linear leg, said upper primaryswitching element of the resonant leg and the bottom primary switchingelement of the linear leg defining a first diagonal of the bridge and,while the first synchronous rectifier is on, transferring power from theprimary side to the secondary side, wherein said transferring ischaracterized by linearly changing, with time, a first amplitude offirst current flowing through the at least one output inductor andlinearly increasing a second amplitude of magnetizing current of saidtransformer to a peak value of the second amplitude; (b) After switchingoff the bottom primary switching element of the linear leg and turningon the upper primary switching element of the linear leg, continuing thetransferring power to the load and continuing the linearly changing ofthe amplitude, of current flowing through the at least one outputinductor, to a lowest value of said first amplitude while maintainingthe second amplitude of the magnetizing current at the peak value; (c)After switching off the upper switching element of the resonant leg,turning on a current-injection switching element corresponding to abottom switching element of the resonant leg with a time delay withrespect to the moment of said switching off the upper switching elementto reflect an injection current, flowing through said current-injectionswitching element, in a secondary winding.
 2. The method according toclaim 1, further comprising: (d) After switching off the upper switchingelement of the resonant leg, discharging a parasitic capacitancereflected across primary switching elements of the resonant leg, withthe use of the leakage inductance; and (e) Switching off the firstsynchronous rectifier after a sum of a magnetizing current, reflected inthe secondary winding, and said injection current reflected into thesecondary winding, exceeds the lowest value of the first amplitude by apredefined excess amount of current to cause the excess amount ofcurrent to flow into at least one primary winding while continuing saiddischarging the parasitic capacitance towards zero.
 3. The methodaccording to claim 2, comprising: (f) Cyclically repeating at leaststeps (a) through (c) with the use of the second synchronized rectifierand a second diagonal of the bridge formed by the upper primaryswitching element of the linear leg and a bottom primary switchingelement of the resonant leg, and the second synchronous rectifier. 4.The method according to claim, 1 further comprising: controlling thesecond amplitude, of the magnetizing current by varying the frequency ofdriving signals of the primary and secondary switching elements.
 5. Themethod according to claim 1, further comprising: varying an amplitude ofthe current injection by varying said time delay.
 6. The methodaccording to claim 1, wherein said operating includes operating the PSFBDC-DC converter the secondary side of which is configured according toany of a) center tap topology, b) a current doubler topology, and c) afull bridge rectifications.
 7. The method according to claim 1, furthercomprising: tailoring an amplitude of the injection current by varyingsaid time delay to cause a discharge of a parasitic capacitance of aprimary switching element of the given leg to zero before said primaryswitching element turns on.
 8. The method according to claim 1,comprising: tailoring the second amplitude of the magnetizing current byvarying a frequency of a driving signal of a switching element from theprimary switching elements to cause a discharge of a parasiticcapacitance of said switching element to zero before said switchingelement turns on.
 9. The method according to claim 1, comprising:tailoring the second amplitude of the magnetizing current by varying afrequency of a driving signal of a switching element from the primaryswitching elements while also tailoring an amplitude of the injectioncurrent injection by varying said time delay to cause a discharge of aparasitic capacitance of said primary switching element to zero beforesaid primary switching element turns on.
 10. A method for operating apulse-shifted full-bridge (PSFB) DC-DC Converter, the convertercomprising a primary side and a secondary side; a transformer having atleast one primary winding at the primary side and at least one secondarywinding at the secondary side, wherein a leakage inductance is formedbetween the at least one primary winding and at least one secondarywinding; a bridge formed by two legs connected in parallel at theprimary side, one leg being a linear leg and another leg being aresonant leg, wherein each leg is formed by corresponding bottom primaryswitching element and upper switching element at the primary sideconfigured in a totem pole arrangement, wherein common terminals of thetwo legs are connected to an input voltage source, wherein sharedterminals of switching elements within one leg, from the two legs, areconnected to one end of at least one primary winding and wherein sharedterminals of the switching elements of another leg, from the two legs,are connected to another end of at least one primary winding, whereinprimary switching elements of a given leg, from the two legs, areconfigured to be complementarty to each other during operation of theconverter with a period of dead time that includes driving signals fromone leg to be phase-shifted with respect to driving signals from anotherleg; first and second synchronous rectifiers at the secondary side; atleast one output inductor at the secondary side, wherein a firstterminal of said at least one output inductor is connected to a load ofthe converter, wherein a second terminal of said at least one outputinductor is directly connected to a synchronized rectifier from thefirst and second synchronous rectifiers; a current-injection electroniccircuit that includes: two current-injection switching elements,respectively corresponding to switching elements in the resonant leg;two current-injection windings disposed on the secondary side andcoupled to the at least one secondary winding of the transformer; andtwo current-injection capacitors; two diodes; and a voltage-injectionvoltage source; wherein said two current-injection switching elementsare connected to respectively-corresponding first terminals of twocurrent-injection windings, wherein each of respectively-correspondingsecond terminals of the two current-injection windings is connected to acorresponding current-injection capacitor from the two current-injectioncapacitors; wherein a cathode of each of the two diodes is connected tosaid corresponding current-injection capacitor at the correspondingsecond terminal and an anode of each of the two diodes is connected tothe voltage-injection voltage source; and a controlling electroniccircuitry configured to generate control signals to the primaryswitching elements, the control signals having square waveforms; themethod comprising: (i) Switching on an upper primary switching elementof the resonant leg and a bottom primary switching element of the linearleg, said upper primary switching element of the resonant leg and thebottom primary switching element of the linear leg defining a firstdiagonal of the bridge and, while the first synchronous rectifier is on,transferring power from the primary side to the secondary side, whereinsaid transferring is characterized by linearly changing, with time, afirst amplitude of first current flowing through the at least one outputinductor and linearly increasing a second amplitude of magnetizingcurrent of said transformer to a peak value of the second amplitude;(ii) After switching off the bottom primary switching element of thelinear leg and turning on the upper primary switching element of thelinear leg, continuing the transferring power to the load and continuingthe linearly changing of the amplitude, of current flowing through theat least one output inductor, to a lowest value of said first amplitudewhile maintaining the second amplitude of the magnetizing current at thepeak value; (iii) After switching off the upper switching element of theresonant leg, discharging a parasitic capacitance reflected acrossprimary switching elements of the resonant leg, with the use of theleakage inductance; (iv) Before switching off the upper switchingelement of the resonant leg, turning on a current-injection switchingelement corresponding to a bottom switching element of the resonant legwith a time interval that is adjustable with respect to the moment ofsaid switching off the upper switching element to reflect an injectioncurrent, flowing through said current-injection switching element, in asecondary winding. (v) Switching off the first synchronous rectifierafter a sum of a magnetizing current, reflected in the secondarywinding, and said injection current reflected into the secondarywinding, exceeds the lowest value of the first amplitude by a predefinedexcess amount of current to cause the excess amount of current to flowinto at least one primary winding while continuing said discharging theparasitic capacitance towards zero.
 11. The method according to claim10, further comprising adjusting said time interval.
 12. The methodaccording to claim 10, comprising: (vi) Cyclically repeating at leaststeps (i) through (III) with the use of the second synchronizedrectifier and a second diagonal of the bridge formed by the upperprimary switching element of the linear leg and a bottom primaryswitching element of the resonant leg, and the second synchronousrectifier.
 13. The method according to claim, 1 further comprising:controlling the second amplitude, of the magnetizing current by varyingthe frequency of driving signals of the primary and secondary switchingelements.
 14. The method according to claim 10, comprising: varying anamplitudes of the injection current by varying said time interval. 15.The method according to claim 10, wherein said operating includesoperating the PSFB DC-DC converter the secondary side of which isconfigured according to any of a) center tap topology, b) a currentdoubler topology, and c) a full bridge rectifications.
 16. The methodaccording to claim 10, comprising: tailoring an amplitude of theinjection current by varying said time delay to cause a discharge of aparasitic capacitance of a primary switching element of the given leg tozero before said primary switching element turns on.
 17. The methodaccording to claim 10, comprising: tailoring the second amplitude of themagnetizing current by varying a frequency of a driving signal of aswitching element from the primary switching elements to cause adischarge of a parasitic capacitance of said switching element to zerobefore said switching element turns on.
 18. The method according toclaim 10, comprising: tailoring the second amplitude of the magnetizingcurrent by varying a frequency of a driving signal of a switchingelement from the primary switching elements while also tailoring anamplitude of the injection current injection by varying said timeinterval to cause a discharge of a parasitic capacitance of said primaryswitching element to zero before said primary switching element turnson.